In recent years, mobile communication becomes widely used, and the number of users who use mobile communication terminals such as portable telephone, automobile telephone, Personal Digital Assistant (PDA) is increasing.
For such mobile communication in Japan, a multiplicity of radio bands are in use: UHF 800 MHz band, or shortly, 800 M band; quasi-microwave band of 1.5 GHz band, or shortly, 1.5 G band; and 1.9 GHz band for use in the PHS (Personal Handy-phone System). Among them, especially in regard to the 800 M band for use in portable telephone services, there is an anxiety of line capacity shortage to come, because the number of users are increasing, and the lines are also used for data communication, et cetera.
Therefore, in order to solve the problem of line capacity shortage, there is a demand for a mobile communication terminal enabling continuation of communication by switching even during talking on a phone to another band having room in line capacity at the moment. For example, the demand for the mobile communication terminal is such that the terminal can deal with both the 800 M band and the 1.5 G band, and that when an applicable channel in the 800 M band becomes short during communication, the connection is automatically hopped to see a channel of the 1.5 G band, or vice versa. Such a mobile communication terminal is referred to as dual-band mobile communication terminal.
That is, the dual-band mobile communication terminal detects, while using one band, an idle slot in another band in real time, to stimulate the base station to shift to the detected idle slot. Thus a shift to the idle slot is performed, which is referred to as MAHO i.e. the mobile assisted hand-over. In addition, the dual-band mobile communication terminal shifts the slot from one band to another in real time based on the instruction of the base station, irrespective of the terminal in waiting of a call or a packet, or already in a communication state.
Such a demand to the dual-band mobile communication terminal also appears in a mobile communication terminal outside the country, because of, for example, commencement of data communication through GPRS (GSM packet radio service) in overseas countries, in which anxiety of frequency shortage to come also has to be taken into consideration.
Further, in IMT-2000, a wideband CDMA service is to be performed using the 2 GHz band, in which a harmonized service with the existent UHF band service is necessary until the service areas for the wideband CDMA are sufficiently expanded. The dual-band mobile communication terminal is also required from this viewpoint.
In order to actualize such a dualband mobile communication terminal, it is necessary for the mobile communication terminal to provide two sets of transmitter/receivers that process respective bands. FIG. 6 shows a block diagram illustrating the configuration of a dual-band mobile communication terminal corresponding to 800 MHz band and 1.5 GHz band, which is structured using conventional technique.
To process the two bands i.e. the 800 M bands and the 1.5 G band, the dual-band mobile communication terminal shown in FIG. 6 is provided with a transmitter/receiver 103, which processes 800 M band signals, and a transmitter/receiver 104, which processes 1.5 G band signals.
In case that a radio frequency signal (RF reception signal) received by an antenna 101 is an 800 M band signal, this RF reception signal is fed to an amplifier 301 provided in transmitter/receiver 103 via an antenna switch 102, and the signal is amplified therein. After the signal amplification, a multiplier (mixer) 303 multiplies the RF reception signal by a local oscillator signal fed from a voltage-controlled oscillator (VCO) 308 to convert the RF reception signal into an intermediate frequency (130 MHz) signal (IF reception signal). Thereafter, the signal is fed to an intermediate frequency section 105.
On the other hand, in case that an RF reception signal is a 1.5 G band signal, this RF reception signal is fed to an amplifier 401 of transmitter/receiver 104 via antenna switch 102, and amplified therein. After the signal is amplified, a multiplier (mixer) 403 multiplies the RF reception signal by a local oscillator signal fed from a voltage controlled oscillator (VCO) 408, to convert the RF reception signal to an IF reception signal having the intermediate frequency (130 MHz), and thereafter, the signal is fed to intermediate frequency section 105.
In intermediate frequency section 105, one of the IF reception signal fed from either transmitter/receiver 103 or transmitter/receiver 104 is selected. After the selected IF reception signal is further frequency-converted, the signal is fed to a baseband processor/controller 106.
Meanwhile, a transmission signal fed to intermediate frequency section 105 from baseband processor/controller 106 is converted to either a signal (IF transmission signal) of 260 MHz in case the transmission signal is to be transmitted through an 800 M band channel, or an IF transmission signal of 82 MHz in case the transmission signal is to be transmitted through a 1.5 G band channel. Thereafter, the signal is fed to either multiplier (mixer) 304 in transmitter/receiver 103 or multiplier (mixer) 404 in transmitter/receiver 104.
In multiplier 304, the IF transmission signal is multiplied by the local oscillator signal fed from VCO 308, and frequency converted into an 800 M-band RF transmission signal. In multiplier 404, the IF transmission signal is multiplied by the local oscillator signal fed from VCO 408, and frequency converted into a 1.5 G-band RF transmission signal. Each RF transmission signal is amplified by an amplifier 302 or 402, and thereafter transmitted from antenna 101 via antenna switch 102.
Here, switchover control of antenna switch 102 is performed by baseband processor/controller 106.
However, in such a dual-band mobile communication terminal, it is not possible to switch over promptly between the channels of the two bands. In the dual-band mobile communication terminal, channel switchover is required in real time, typically within an allowed limit of approximately 1 millisecond (ms) for the switchover. In contrast, according to the configuration shown in FIG. 6, the switchover time from the 800 M band to the 1.5 G band is on the order of 4 ms. The reason for this is shown below.
In both the 800 M band and the 1.5 G band, the RF reception signal and the RF transmission signal have variable frequencies, which vary within a predetermined range (810 MHz–885 MHz in case of the 800 M-band RF reception signal, as shown later in FIG. 1) in accordance with the channel allocated at the time of communication.
In order to frequency-convert the RF reception signal of variable frequency into a constant IF reception signal of 130 MHz, or to frequency-convert a constant IF transmission signal of either 260 MHz or 82 MHz to the RF transmission signal of variable frequency, the frequency of the local oscillator signal output from VCO 308 or 408 is controlled to vary in accordance with the RF reception signal frequency or the RF transmission signal frequency.
Now, because it becomes complicated to describe these frequencies using a variable range, a representative value is defined for explanation purpose. Based on this representative value, the switchover time requiring 4 ms is explained in the following.
A representative value of an RF transmission frequency of the 800 M band is set to 949 MHz (the center value of the transmission frequency, i.e. the uplink frequency), and a representative value of a reception frequency is set to 819 MHz (the center value of the reception frequency, i.e. the downlink frequency). Further, a representative value of a local oscillator frequency is set to 689 MHz (which is the local oscillator frequency in case of the transmission frequency of 949 MHz and the reception frequency of 819 MHz). Also, a representative value of an RF transmission frequency of the 1.5 G band is set to 1,441 MHz (the center value of the uplink frequency), and a representative value of a reception frequency is set to 1,489 MHz (the center value of the downlink frequency). Further, a representative value of a local oscillator frequency is set to 1,359 MHz (which is the local oscillator frequency in case of the transmission frequency of 1,441 MHz and the reception frequency of 1,489 MHz).
The local oscillator signal of the 800 M band is generated by a phase lock loop (PLL) structured of a signal loop originating from an integer divider 305, passing through a phase comparator 306, a low pass filter (LPF) 307, and VCO 308, and returning to integer divider 305. Similarly, the local oscillator signal of the 1.5 G band is generated by a PLL structured of a signal loop originating from an integer divider 405, passing through a phase comparator 406, an LPF 407, and VCO 408, and returning to integer divider 405.
A comparison frequency input to phase comparators 306 and 406 is set to no greater than 25 kHz, so as to synthesize the local oscillator frequencies at 25 kHz intervals, which are channel intervals in both the 800 M band and the 1.5 G band. Here, the comparison frequency is set to 25 kHz, as shown in FIG. 6.
In this case, at the local oscillator frequency of 1,359 MHz of the 1.5 G band, the frequency division number N of integer divider 405 becomes: N=1,359 MHz/25 kHz=54,360.
The loop gain of the PLL is inversely proportional to the frequency division number N. Therefore, the PLL loop gain becomes smaller in the 1.5 G band. This is the reason of the channel switchover time becoming approximately 4 ms.
More specifically, as LPF 307 and LPF 407 in the mobile communication terminal, a filter, referred to as a lag-lead filter, of quadratic delay including fixed resistors R1, R2 and a fixed capacitor C is generally employed, as shown in FIG. 7. This filter circuit is of simple structure enabling a miniaturized LPF, and further a natural angular frequency ωn and a dumping factor ζ can be set independently. These are the reasons for use of such a filter.
According to the LPF shown in FIG. 7, as shown in the calculation result in FIG. 7, natural angular frequency becomes: ωn=2.266K [rad/s] (360.8 Hz), and the PLL converges with vibrating with a natural vibration period T=1/Fn=1/360.8=2.772 ms. Here, Fn is a natural frequency, and is represented by Fn=ωn/2π.
The transitional response thereof is that the vibration almost converges in approximately three times of the bound (rebound), as shown in FIG. 8, on assumption of the dumping factor ζ=0.6 (where, 0.5–0.7 is considered appropriate). Here, a range to be regarded as converged is that a phase error falls within the range of ±π/10 (a local oscillator frequency error falls within approximately ±1 kHz). The time consumed until the phase error (or frequency error) falls within the range regarded as converged is considered as a convergence time.
Accordingly, when the switchover is performed from the 800 M band to the 1.5 G band, the time consumed from the response to the convergence is approximately 1.5 times as long as the vibration period T=2.772 ms, that is, approximately 4 ms. In other words, a time approximating 4 ms is necessary from the beginning of the switchover from the 800 M band to the 1.5 G band until the local oscillator frequency comes to a proper frequency corresponding to the reception channel or the transmission channel. Meanwhile, with regard to the switchover from the 1.5 G band to the 800 M band, the transition time becomes approximately 2 ms, as a result of calculation similar to the above.
In contrast, there is another way to synthesize a local oscillator signal by use of a fractional divider to achieve a high speed switchover, as shown in FIG. 9. The configuration shown in FIG. 9 is the same as the configuration shown in FIG. 6, except for integer dividers 305 and 405 shown in FIG. 6 are respectively replaced by fractional dividers 315 and 415 in FIG. 9, and further the comparison frequencies are replaced to 400 kHz and 600 kHz.
Fractional dividers 315, 415 can take a fractional number as the frequency division number (for example, the frequency division number=54,383/24=2,265+23/24=2,265.9583 (circulating decimal)), while the integer divider has an integer frequency division number only.
An advantage of the fractional divider lies in that the intervals of the frequency division number less than an integer value (for example, at intervals of 0.1) is settable. With this, the comparison frequency higher than 25 kHz is settable. This enables a larger PLL loop gain K, and as a result, the convergence time can be shortened. For example, assuming the local oscillator frequency is 1,359,575 MHz, and the frequency division number is 54,383/24=2,265+23/24=2,265.95833 . . . (circular decimal), the comparison frequency becomes 1,359.575 MHz/{2,265+(23/24)}=6.00 kHz.
This figure corresponds to 24 times as much as the comparison frequency shown in FIG. 6, (24=600 kHz/25 kHz), and the PLL loop gain K multiplied by this multiple number can be obtained. Accordingly, the frequency convergence time becomes approximately 1/24 of that shown in FIG. 6, and as a result, the frequency convergence time=4 ms*( 1/24)≈166.7 microsecond (μs) can be obtained. Thus, it becomes possible to satisfy the requirement for the dual-band mobile communication terminal.
However, when using such a fractional divider, an irregular jitter caused by the fraction adjustment of the frequency division number is produced in the divided signal waveform. This is equivalent to superposed noise, producing noise superposed in the synthesized local oscillator frequency. As a result, there arises a problem of deteriorated spectral purity produced in the local oscillator signal, as shown in FIG. 10.
This deterioration of the signal spectral purity in the local oscillator frequency produces another problem in the mobile communication terminal: a problem of difficulty in conforming to the standards of adjacent channel leakage power in the transmission characteristics of the mobile communication terminal, and also to the standard of adjacent channel selectivity and next adjacent channel selectivity in the reception characteristics, as well as the anti-interference capability represented by intermodulation response rejection.
Further, the fractional divider and the phase comparator is desirably structured of a single monolithic integrated circuit, so as to achieve a miniaturized and lightweight mobile communication terminal. However, cost for structuring the fractional divider into the integrated circuit is high. Namely, a large circuit scale makes it difficult to produce an integrated circuit including a frequency divider operational at a frequency over 1 GHz, in which an arbitrary fractional frequency division number is settable. Additionally, technique of introducing an addition circuit has been proposed, so as to overcome the above-mentioned problem related to the fraction processing against the fractional frequency division number. However, because of this addition circuit, the circuit scale becomes larger. Manufacturing a monolithic LSI for the PLL also brings about higher cost, which retrogrades against technological trend to manufacture a low-cost mobile communication terminal.